Variable inductor, oscillator including the variable inductor and radio terminal comprising this oscillator, and amplifier including the variable inductor and radio terminal comprising this amplifier

ABSTRACT

An amplifier comprises an amplifier circuit which comprises a first inductor as an impedance element for degeneration, and a control circuit which has a second inductor electro-magnetically connected to the first inductor, and changes a control current flowing through the second inductor to change an inductance value of the first inductor, thereby changing amplification characteristics of the amplifier circuit.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is based upon and claims the benefit of priority fromthe prior Japanese Patent Applications No. 2002-160621, filed May 31,2002; No. 2002-188946, filed Jun. 28, 2002; and No. 2002-270984, filedSep. 18, 2002, the entire contents of all of which are incorporatedherein by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates to a variable inductor, and more particularly toa variable inductor which uses a circuit constituted of an activeelement and a plurality of interconnected inductors to changeinductance.

Additionally, this invention relates to an oscillator having aninductor, and a radio terminal having an inductor, and also relates toimprovements in a circuit designing technology of an amplifier and aradio terminal and a gain varying method of the amplifier.

2. Description of the Related Art

Generally, in order to vary characteristics of an electronic circuit,characteristics of an active element or a value of a passive elementincluded in the circuit is changed. For the active element, thecharacteristics of the active element can be changed by changing a biasvoltage applied to the active element. For the passive element, apassive element, for example, a variable resistive element can be easilyrealized by using ON resistance of a MOSFET, and a variable capacitiveelement can be easily realized by using a pn junction.

For an inductor as a passive element, it is generally considereddifficult to vary inductance while maintaining good characteristics. Amethod which uses an active element to constitute an inductor and variesinductance is disclosed in ELECTRONICS LETTERS 2nd January, Vol. 28, No.21, pp. 78 to 80, 1992. However, there is a problem of badcharacteristics of noise or distortion because of use of the activeelement for the inductor.

Thus, various technologies have been presented, which vary inductancewithout constituting an inductor of an active element. However, all ofthese technologies have problems in practical application. For example,Jpn. Pat. Appln. KOKAI Publication No. 8-162331 discloses a method whichinserts a switch into the middle of an inductor, and turns the switchON/OFF to change inductance. This method has a problem that ONresistance of the switch deteriorates performance of the variableinductor.

Jpn. Pat. Appln. KOKAI Publication No. 2000-223317 discloses a methodwhich physically changes a shape of an inductor by a laser beam. Thismethod necessitates a physical adjustment of the inductor after it ismanufactured. Thus, there are problems of high manufacturing costs and adifficulty of changing inductance in a situation where a circuit isoperated.

Jpn. Pat. Appln. KOKAI Publication No. 7-320942 discloses a technologywhich constitutes a variable inductor by using interconnection of aplurality of inductors. According to this method, a shape of theinductor is physically changed in order to change a couplingcoefficient. Thus, there are problems of miniaturization and low costsof a circuit which constitutes the variable inductor.

Further, D. R. Pehlke et al. have presented, in U.S. Pat. No. 5,994,985,a technology which uses a directional coupler to separate an inputsignal into two, and controls amplitudes and phases of signals flowingthrough two interconnected inductors to change inductance. However,since the directional coupler is not generally suited for integration,there is a problem that it is difficult to realize a variable inductorby an integrated circuit.

By realizing the variable inductor, it is also possible to control evena voltage-controlled oscillator (VOC) which includes an LC resonantcircuit (inductor: inductance L, capacitor: capacity C). An oscillationfrequency (f) of the voltage controlled oscillator which includes thisLC resonant circuit is generally represented by f=½[2π(LC)½]. If theinductance l or the capacity C is controlled, the oscillation frequency(f) is controlled. However, in the conventional LC resonant circuit, itis difficult to realize a variable inductor as described above.Generally, therefore, the capacity C is varied, for example, a reversebias application voltage to a pn junction diode is changed to vary thecapacity C and, by changing this capacity, the oscillation frequency ischanged.

If such a voltage controlled oscillator is formed as an integratedcircuit on a semiconductor substrate, i.e., IC formation, a parasiticcapacitance, e.g., a parasitic capacitance of the inductor, a drainparasitic capacitance of a MOS transistor, a gate parasitic capacitanceof the MOS transistor or the like, is generated. Generation of such aparasitic capacitance is inevitable, and thus there is a problem thatsuch a parasitic capacitance reduces a fluctuation width in a variablecapacity C of the LC resonant circuit. For example, if a fluctuationamount of the capacity C is AC, a parasitic capacitance amount is addedto a denominator as a non-fluctuation portion while a design fluctuationrate is assumed to be AC/C. Thus, in practice, there is a problem thatthe fluctuation rate becomes small, i.e., ΔC/(C+parasitic capacitance).Though it is dependent on a circuit design or the like, if the capacityC and the parasitic capacitance are about equal, a change rate isreduced to about ½.

Because of the presence of such a parasitic capacitance, a ratio of thecapacity value of the LC resonant circuit occupied by the variablecapacity C is inevitably reduced. Thus, compared with a change rate ofthe variable capacity C, a change in the capacity value of the LCresonant circuit is reduced and, consequently, a variable range of anoscillation frequency is narrowed.

Today, however, a frequency band used for a portable telephone, a radioLAN device or the like has been widened, and there is a case where aplurality of frequency bands are dealt with by one device. Thus, therehas been an increase in demand for expansion of a change width of theoscillation frequency. From this viewpoint, realization of a variableinductor is desired.

The variable inductor can be applied to an amplifier provided with aninductor. For example, in an amplifier which comprises the inductor fordegeneration, if its inductance is reduced, distortion characteristicsare deteriorated while a gain and noise characteristics of the amplifierare improved. Conversely, if the inductance is increased, a gain andnoise characteristics are deteriorated while distortion characteristicsare improved. Because of this trade-off relation, an inductance value isdecided to obtain desired characteristics when the amplifier isdesigned.

Generally, realization of low distortion characteristics whilemaintaining a high gain and low noise characteristics is dealt with byincreasing the amount of a current. In the amplifier used for a receiverof a radio terminal, characteristics necessary for the amplifier aredifferent depending on a sized of a received signal. Generally, since itis considered important to amplify a signal at low noise if the receivedsignal is small, a good gain and good noise characteristics are requiredof the amplifier. On the other hand, if the received signal is large,good distortion characteristics are required.

In the conventional amplifier which comprises the inductor fordegeneration, since inductance is fixed, the amount of a suppliedcurrent is controlled in order to change the characteristics of theamplifier. That is, to improve distortion characteristics, a current iscontrolled so as to increase the amount of a supplied current. However,an increase in the amount of a current made to change thecharacteristics of the amplifier creates a problem of increased powerconsumption.

Further, there is disclosed a circuit example of a variable gainamplifier in Dual-Band High-linearity Variable-Gain Low-Noise Amplifierfor Wireless Applications K. L. Fong, “Dual-Band High-LinearityVariable-Gain Low-Noise Amplifier for Wireless Applications,” IEEEISSCC99, pp 224 to 225, 1999. In this variable gain amplifier, afirst-stage common emitter circuit constituted of a first transistor Q1is always operated, and gain switching is realized by switching secondto fourth transistors Q2 to Q4 which constitute a first-stage commonbase circuit. Since the first transistor Q1 is operated, input impedanceis not greatly changed even if a gain is switched. However, a fixedcurrent is always consumed, and distortion characteristics aresubstantially constant.

However, in the conventional circuit disclosed in Dual-BandHigh-Linearity Variable-Gain Low-Noise Amplifier for WirelessApplications K. L. Fong, “Dual-Band High-Linearity Variable-GainLow-Noise Amplifier for Wireless Applications,” IEEE ISSCC99, pp 224 to225, 1999, there is a problem that a large current is consumed even whena gain is low, and distortion characteristics are about similar to thosewhen the gain is high. To realize an amplification stage of a basicallyhigh gain and good distortion characteristics, current consumption isnecessary to a certain extent. In the case of a gain which is not sohigh or to attenuate a signal, it is possible to realize anamplification stage of good distortion characteristics without consuminga current so greatly. However, if a plurality of amplification stagesare switched, there is a problem of a change in input impedance.

As described above, the variable inductor of the conventional art has aproblem in electric characteristics, and there is a problem that it isdifficult to realize miniaturization, low costs and an integratedcircuit.

In the radio terminal such as a portable telephone, there is a strongdemand for making an adaptive characteristic change of amplifiercharacteristics in accordance with a received signal level, while lowerpower consumption is similarly demanded strongly. In the amplifier whichuses a fixed inductor for degeneration, the only way to improvedistortion characteristics is to increase the amount of a current, whichbrings about an increase in power consumption.

BRIEF SUMMARY OF THE INVENTION

An object of the present invention is to provide a variable inductorwhich has good electric characteristics, allows easy miniaturization andlow costs, and is suited to circuit integration.

According to an aspect of the present invention, there is provided avariable inductor comprising:

a signal input terminal which receives an input signal;

a distributor including first and second active elements, configured tovary a distribution ratio of first and second currents flowing throughthe first and second active elements, respectively;

output terminals which output the first and second currents,respectively; and

a first inductor through which the first current flows;

a second inductor through which the second current flows and which isconnected to the first inductor; and

According to an another aspect of the present invention, there is alsoprovided an oscillator comprising:

a voltage control oscillation circuit having a first inductor; and

a frequency control circuit having a second inductorelectro-magnetically coupled to the first inductor, configured to supplya control current to the second inductor, and controls an oscillationfrequency of the voltage control oscillation circuit by changing thecontrol current to change an inductance value of the first inductor.

According to an yet another aspect of the present invention, there is anamplifier comprising:

an amplifier circuit having a first inductor; and

a control section having a second inductor electro-magnetically coupledto the first inductor, configured to supply a control current to thesecond inductor, and change the control current to change an inductancevalue of the first inductor, thereby changing amplificationcharacteristics of the amplifier circuit.

According to an further aspect of the present invention, there is anamplifier comprising:

an input terminal which receives an input signal;

a variable gain amplification circuit configured to amplify the inputsignal with a variable gain of an amplification, which includes aplurality of amplification stages arranged in parallel; and

an input impedance adjustment circuit including a variable resistorcircuit connected to the input terminal, configured to adjust aresistance value of the variable resistor to compensate a change ininput impedance in accordance with a change of the gain.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWING

FIG. 1 is a circuit diagram schematically showing a variable inductoraccording to a first embodiment of the present invention.

FIG. 2 is a circuit diagram schematically showing a modified example ofthe variable inductor shown in FIG. 1.

FIG. 3 is a circuit diagram schematically showing a variable inductoraccording to a second embodiment of the present invention.

FIG. 4 is a circuit diagram schematically showing a modified example ofthe variable inductor shown in FIG. 3.

FIG. 5 is a circuit diagram schematically showing a variable inductoraccording to a third embodiment of the present invention.

FIG. 6 is a circuit diagram schematically showing a variable inductoraccording to a fourth embodiment of the present invention.

FIG. 7 is a circuit diagram schematically showing a variable inductoraccording to a fifth embodiment of the present invention.

FIG. 8 is a circuit diagram schematically showing a variable inductoraccording to a sixth embodiment of the present invention.

FIG. 9 is a circuit diagram schematically showing a variable inductoraccording to a seventh embodiment of the present invention.

FIG. 10 is a circuit diagram schematically showing a variable inductoraccording to a modification of a seventh embodiment of the presentinvention.

FIG. 11 is a block diagram schematically showing an oscillator accordingto an embodiment of the present invention.

FIG. 12 is a block diagram schematically showing an oscillator accordingto another embodiment of the present invention.

FIG. 13 is a block diagram schematically showing a circuit where avoltage-current conversion circuit of FIG. 12 is constituted ofcascode-connected transistors.

FIG. 14 is a block diagram schematically showing an oscillator accordingto a yet another embodiment of the present invention where a pluralityof differential pairs are arranged.

FIG. 15 is a block diagram schematically showing an oscillator accordingto a still another embodiment of the present invention which uses avariable phase shifter and a variable gain amplifier.

FIG. 16 is a block diagram schematically showing Colpitts oscillationcircuit according to a yet another embodiment of the present invention.

FIG. 17 is a plan view schematically showing a layout pattern of theinductor.

FIG. 18 is a plan view schematically showing another layout pattern ofthe inductor.

FIG. 19 is a plan view schematically showing yet another layout patternof the inductor. Schematic view of the inductor explaining theembodiment of the present invention.

FIG. 20 is a block diagram schematically showing a radio terminal whichcomprises the oscillator shown in each of FIGS. 11 to 16.

FIG. 21 is a block diagram schematically showing an amplifier accordingto an embodiment of the present invention.

FIG. 22 is a circuit diagram showing a specific circuit of thedifferential amplifier shown in FIG. 21.

FIG. 23 is a block diagram showing an amplifier according to an anotherembodiment of the present invention.

FIG. 24 is a circuit diagram showing a specific circuit of thedifferential amplifier shown in FIG. 23.

FIG. 25 is a circuit diagram showing another specific circuit of thedifferential amplifier shown in FIG. 23.

FIGS. 26A and 26B are views each showing a circuit example of a variableresistor shown in FIG. 25.

FIG. 27 is an equivalent circuit diagram showing an input impedance ofamplification stages shown in FIGS. 24 and 25.

FIG. 28 is a circuit diagram showing a circuit of a modified example ofa circuit shown in FIG. 25.

FIG. 29 is a circuit diagram showing yet another specific circuitexample of the differential amplifier shown in FIG. 21.

FIG. 30 is a circuit diagram showing yet another circuit of a modifiedexample of the differential amplifier shown in FIG. 23.

FIG. 31 is a block diagram schematically showing a radio terminal whichincorporates the amplifier described above with reference to each ofFIGS. 21 to 30.

FIG. 32 is a block diagram showing basic circuitry of an amplifier whichcomprises an inductor having no mutual inductance according to ananother embodiment of the present invention.

FIG. 33 is a block diagram showing a circuit to realize a variableresistor shown in FIG. 32.

FIG. 34 is a block diagram showing a yet another circuit to realize thevariable resistor shown in FIG. 32.

FIG. 35 is a block diagram showing a circuit to realize the variableresistor shown in FIG. 34.

FIG. 36 is a graph showing a result of simulation of input impedance ofa portion excluding a MOSFET of an input portion in the circuit shown inFIG. 35.

FIG. 37 is a block diagram showing a yet another circuit to realize thevariable resistor shown in FIG. 34.

FIG. 38 is a block diagram showing a still another circuit to realizethe variable resistor shown in FIG. 34.

FIG. 39 is a block diagram showing a circuit example of a radio terminalwhere an amplifier circuit of FIG. 32 is applied to a low-noiseamplifier of a radio equipment.

DETAILED DESCRIPTION OF THE INVENTION

Next, detailed description will be made of a variable inductor, anoscillator which incorporates the variable inductor in a circuit, aradio terminal which comprises this oscillator, an amplifier whichincorporates the variable inductor in a circuit, and a radio terminalwhich comprises this amplifier according to the embodiments of thepresent invention with reference to the accompanying drawings.

The description will be made by way of example where an FET is used as atransistor which is an active element used for a distributor of avariable inductor circuit. However, a bipolar transistor can be used inplace of the FET to realize the variable inductor circuit.

<Variable Inductor>

First Embodiment

First, description will be made of a variable inductor according to abasic embodiment of the present invention. FIGS. 1 and 2 show circuitryof a viable inductor of a first embodiment of the present invention. Aninput signal (Input) input to a signal input terminal 11 is distributedto a plurality of signal paths 13 a, 13 b, . . . , 13 n by a distributor12 constituted by using an active element. Inductors 14 a, 14 b, . . .14 b are inserted into the signal paths 13 a, 13 b, . . . , 13 n. Theinductors 14 a, 14 b, . . . 14 n are constituted of, e.g., spirallyformed leads, and arranged close to one another to be interconnected.

The inductors 14 a, 14 b, . . . 14 n generate magnetic fluxes dependingon sizes of input signals, and each magnetic flux is applied on theother close inductor to interconnect each inductor with this otherinductor. Thus, the inductors 14 a, 14 b, . . . 14 n have selfinductance caused by a magnetic flux generated by each inductor itself,and mutual inductance decided by a magnetic flux generated by the otherinductor coupled with each inductor. For example, inductance of theinductor 14 a, i.e., inductance between terminals 15, 16 (calledvariable inductor terminals) of both ends of the inductor 4 a is decidedby self inductance Lsa of the inductor 14 a, and mutual inductance Mab,. . . , Man between the inductor 14 a and the other inductors 14 b, . .. 14 n. Here, depending on directions of the interconnected inductors(direction considering both of current and winding wire directions), theinterconnected inductors can be set in a relation of reinforcing orweakening magnetic fluxes generated by the inductors. Thus, theinductance La between the variable inductor terminals 15, 16 can be setlarge or small with respect to the self-inductance Lsa.

The signals distributed from the distributor 12 to the signal paths 13a, 13 b, . . . , 13 n are supplied to the inductors 14 a, 14 b, . . . 14n. Here, by adjusting a distribution ratio of the signals distributedfrom the distributor 12 to the inductors 14 a, 14 b, . . . 14 n, themutual inductance Mab, . . . , Man, i.e., the amount of a magnetic fluxby interconnection of the inductors 14 a, 14 b, . . . 14 n, can becontrolled. Thus, the inductance between the variable inductor terminals15, 16 can be set to a desired value.

The distributor 12 is constituted of an active element such as atransistor as described later. Accordingly, different from theconventional variable inductor circuit which uses the directionalcoupler, the distributor 12 shown in FIGS. 1 and 2 can be easily formedinto an integrated circuit. In the circuit shown in FIG. 1, thedistributor 12 is connected between the signal input terminal 11 and oneend of each of the inductors 14 a, 14 b, . . . 14 n. In the circuitshown in FIG. 2, one end and the other end of each of the inductors 14a, 14 b, . . . 14 n are connected to the distributor 12.

Next, some more specific embodiments of the variable inductor shown inFIGS. 1 and 2 will be described.

Second Embodiment

FIGS. 3 and 4 show a variable inductor where the distributor 12 includesa common source circuit according to the embodiment. The circuit of thisembodiment is equivalent to the specific circuit example of the basiccircuitry shown in FIG. 1. The input signal from the signal inputterminal 11 is amplified by a common source circuit, e.g., MOSFET's(simply referred to as transistor, hereinafter) 21 a, 21 b to bedistributed to two signal paths, and supplied to the inductors 14 a, 14b inserted into these signal paths. Gate terminals of the transistors 21a, 21 b are connected to the signal input terminal 11, source terminalsare connected to ends of the inductors 14 a, 14 b, and drain terminalsare connected to output terminals 22 a, 22 b of the variable inductor.In this variable inductor, signal currents amplified by the MOSFET's 21a, 21 b are outputted from output terminals 22 a, 22 b of the variableinductor.

The other ends of the inductors 14 a, 14 b are connected to ends ofpower sources 23 a, 23 b and ends of capacitors 24 a, 24 b, and groundedwith respect to AC components (high frequency components). The otherends of the power sources 23 a, 23 b and the other ends of thecapacitors 24 a, 24 b are connected to the ground. Among currentsflowing through the inductors 14 a, 14 b, DC components flow through thepower sources 23 a, 23 b, and AC components (high frequency components)are bypassed by the capacitors 24 a, 24 b.

In the variable inductor shown in FIGS. 3 and 4, black points given nearsymbols indicating the inductors 14 a, 14 b represent winding startpositions of the inductors. In the inductors where these positions arethe same, directions of all the winding wires thereof are the same, andphases of magnetic fluxes are the same.

If the directions of the inductors 14 a, 14 b are the same as shown inFIG. 3, magnetic flues are generated in reinforcing directions in theinductors 14 a, 14 b. Thus, if self inductance of the single inductor 14a is Lsa, self inductance of the single inductor 14 b is Lsb, and ancoupling coefficient between the inductors 14 a, 14 b is kab, effectiveinductance of the inductor 14 a considering interconnection, i.e.,inductance La between the variable inductor terminals 15, 16 isrepresented by the following equation (1):La=lsa+kab·Lsb   (1)

If mutual inductance between the inductors 14 a, 14 b is Mab, theequation (1) is represented by the following equation (2):La=Lsa+Mab   (2)

On the other hand, if the directions of the inductors 14 a, 14 b areopposite as shown in FIG. 4, magnetic fluxes are generated in aweakening relation in the inductors 14 a, 14 b. This inductance La isrepresented by the following equation (3) or (4):La=Lsa−kab·Lsb   (3)La=Lsa−Mab   (4)

Here, the coupling coefficient kab, i.e., mutual inductance Mab, isdecided based on a physical arrangement of the inductors 14 a, 14 b,sizes of the transistors 21 a, 21 b, current values Ia, Ib or the powersources 23 a, 23 b etc. Thus, even without changing the physicalarrangement of the inductors 14 a, 14 b, by adjusting the current valuesIa, Ib or the sizes of the transistors 21 a, 21 b, a value of theinductance La can be varied.

For example, the current source 23 b is set as a variable current sourcewhich can control the current value Ib by a control signal from theoutside, and this current value Ib is continuously changed. Accordingly,the mutual inductance Mab is changed. Thus, the inductance La betweenthe variable inductor terminals 15, 16 is continuously changed. When thepower source 23 b is turned ON/OFF, the inductance La can be switchedbetween (Lsa+Mab) or (Lsa−Mab) and Lsa in a binary manner.

According to the embodiment shown in FIGS. 3 and 4, the common sourcecircuit by the FET is used for the distributor 12. However, a commonemitter circuit by a bipolar transistor can also be used. In the case ofthe emitter-grounded circuit, the gate terminal, the drain terminal andthe source terminal of the FED are replaced by a base terminal, acollector terminal and an emitter terminal of the bipolar transistor. Inthe circuit shown in FIGS. 1 and 2, the two inductors are used. However,even for a circuit which uses three or more inductors, a constitutionsimilar to the embodiment can be applied.

Moreover, as a modified example, the distributor 12 may be constitutedof a plurality of transistors where gate or base terminals ofdistributors are connected in common to the signal input terminal 11through at least one inductor of the plurality of inductors. As anothermodified example of the embodiment, the distributor 12 may beconstituted of a plurality of transistors where gate or base terminalsare connected in common to the signal input terminal 11, and drain orcollector terminals are connected to ends of the plurality of inductors.

Third Embodiment

FIG. 5 shows a variable inductor according to a third embodiment whichuses a transistor circuit where a gate is grounded to a distributor 12.A source terminal of at least one of first transistors 31 a, 31 b, . . ., 31 n, and source terminals of a plurality of second transistors 32 a,32 b, . . . , 32 n are connected to a signal input terminal 11. Drainterminals of the first transistors 31 a, 31 b, . . . , 31 n areconnected to one end of a first inductor 14 a, and gate terminals areconnected to control signal input terminals 33 a, 33 b, . . . , 33 n.Drain terminals of the second transistors 32 a, 32 b, . . . , 32 c areconnected in common to one end of a second inductor 14 b, and gateterminals are connected to control signal input terminals 34 a, 34 b, .. . , 34 n.

Control signals φ33 a, φ33 b, . . . , φ33 n are input to the controlsignal input terminals 33 a, 33 b, . . . , 33 c. Control signals φ34 a,φ34 b, . . . , φ34 n are input to the control signal input terminals 34a, 34 b, . . . , 34 c. When the control signals φ33 a, φ33 b, . . . ,φ33 n and φ34 a, φ34 b, . . . , φ34 b are changed in a binary manner, adistribution ratio of signal levels to the inductors 14 a, 14 b, i.e., aratio of currents flowing through the inductors 14 a, 14 b, is changed.Thus, mutual inductance Mab between the inductors 14 a, 14 b is changedand, as a result, effective inductance La (inductance between terminals15, 16) of the inductor 14 a can be changed.

A distribution ratio of a signal level to the inductor 14 a is decidedbased on the number of transistors turned ON by the control signals φ33a, φ33 b, . . . , φ33 n among the transistors 31 a, 31 b, . . . , 31 c.Similarly, a distribution ratio of a signal level to the inductor 14 bis decided based on the number of transistors turned ON by the controlsignals φ34 a, φ34 b, . . . , φ34 n among the transistors 32 a, 32 b, .. . , 32 c.

The inductance La between the terminals 15, 16 may be continuouslychanged by setting the control signals φ33 a, φ33 b, . . . , φ33 n, andφ34 a, φ34 b, . . . , φ34 n as analog signals and continuously changingcurrents flowing through the transistors 31 a, 31 b, . . . , 31 n andthe transistors 32 a, 32 b, . . . , 32 c.

In FIG. 5, the plurality of transistors 31 a, 31 b, . . . , 31 n areconnected to the inductor 14 a. However, if a small variable range ofinductance is allowed, the number of first transistors may be one.Similarly, the plurality of second transistors 32 a, 32 b, . . . , 32 nare connected to the inductor 14 b. However, if a small variable rangeof inductance is allowed, the number of second inductors may be one.

In the variable inductor of the embodiment, the common gate circuit bythe FET is used for the distributor 12. However, a common base circuitby a bipolar transistor can also be used. In the case of thebase-grounded circuit, the gate terminal, the drain terminal and thesource terminal of the FED are replaced by a base terminal, a collectorterminal and an emitter terminal of the bipolar transistor.

Moreover, as a modified example, the distributor 12 may be constitutedof at least one first transistor where a source or emitter terminal isconnected to the signal input terminal 11 through at least one firstinductor, and a gate or base terminal is connected to the control signalinput terminal, and at least one second transistor where a source oremitter terminal is connected to the signal input terminal 11 throughthe first inductor, a drain or collector terminal is connected to oneend of at least one second inductor coupled to the first inductor, and agate or base terminal is connected to the control signal input terminal.

Fourth Embodiment

FIG. 6 shows a variable inductor according to a fourth embodiment of thepresent invention which uses a source follower circuit for a distributor12. Gate terminals of a plurality of transistors 41 a, 41 b (two in theexample of FIG. 6) are connected to a signal input terminal 11. Drainterminals of the transistors 41 a, 41 b are connected to a power sourceVdd which is a constant potential point, and source terminals areconnected to ends of inductors 14 a, 14 b. Further, current sources 43a, 43 b are connected to the source terminals of the transistors 41 a,41 b. Thus, the transistors 41 a, 41 b operate as source followercircuits.

Here, as in the case of the second embodiment, if the current source 43a is set as a variable current source which can control a current valueIb by a control signal from the outside, and this current value Ib iscontinuously changed, mutual inductance Mab between the inductors 14 a,14 b is accordingly changed, whereby inductance La between the terminals15, 16 is continuously changed. When the current source 43 b is turnedON/OFF, the inductance La is switched between (Lsa+Mab) or (Lsa−Mab) andLsa in a binary manner where self-inductance of the inductor 14 a is setas Lsa.

According to the embodiment, the source follower circuit by the FET isused for the distributor 12. However, it is apparent that an emitterfollower circuit by a bipolar transistor can also be used.

Further, as a modified example of the embodiment, the distributor 12 maybe constituted of a plurality of transistors where gate or baseterminals are connected to the signal input terminals 11 through theinductors, and the drain or collector terminals are connected to theconstant potential point.

Fifth Embodiment

FIG. 7 shows a variable inductor of a fourth embodiment of the presentinvention which uses cascode-connected circuits for a distributor 12.Gate terminals of first and second transistors 51, 52 are connected to asignal input terminal 11. Source terminals of the transistors 51, 52 areconnected to current sources 53, 54. A drain terminal of the firsttransistor 51 is connected to a source terminal of a third transistor55, and a drain terminal of the second transistor 52 is connected incommon to source terminals of a plurality of fourth transistors 56 a, 56b (two in the example of FIG. 7). That is, the transistor 51 and thetransistor 55 are cascode-connected, and the transistor 52 and thetransistors 56 a, 56 b are cascode-connected.

Drain terminals of the third transistor 55 and the fourth transistors 56a, 56 b are connected to ends of inductors 14 a, 14 b, 14 c. A gateterminal of the third transistor 55 is connected to a control signalinput terminal 58, and gate terminals of the fourth transistors 56 a, 56b are connected to control signal input terminals 57 a, 57 b. Asindicated by black points given near symbols of the inductors 14 a, 14b, 14 c, the inductor 14 b is arranged in a magnetic flux reinforcingdirection, and the inductor 14 c is arranged in a magnetic fluxweakening direction with respect to the inductor 14 a.

Now, in a state where a control signal φ is input to the control signalinput terminal 58 to turn ON the transistor 55, if a control signal φ+is input to the control signal input terminal 57 a and a control signalφ− is input to the control signal input terminal 57 b to turn ON thetransistor 56 a connected to the inductor 14 b and turn OFF thetransistor 56 b connected to the inductor 14 c, inductance La betweenboth ends 15, 16 of the inductor 14 a becomes larger than selfinductance Lsa of the inductor 14 a. Conversely, if a control signal φ−is input to the control signal input terminal 57 a and a control signalφ+ is input to the control signal input terminal 57 b to turn OFF thetransistor 56 a and turn ON the transistor 56 b, inductance La betweenboth ends 15, 16 of the inductor 14 a becomes smaller compared with selfinductance Lsa.

Thus, by using the cascode-connected circuits for the distributor 12,the fourth transistors 56 a, 56 b cascode-connected to the secondtransistor 52 are turned ON/OFF to selectively supply signals to theinductors 14 b, 14 c. As a result, the inductance La between theterminals 15, 16 can be increased/decreased.

According to the embodiment, the cascode-connected circuits by the FETare used for the distributor 12. However, it is apparent thatcascode-connected circuits by a bipolar transistor can also be used.

Sixth Embodiment

FIG. 8 shows a variable inductor according to a fifth embodiment of thepresent invention. A gate terminal of a transistor 61 which constitutesan amplifier 60 is connected to a signal input terminal 11. A currentsource 63 and an AC (high frequency) bypass capacitor 64 are connectedto a source terminal of the transistor 61. One end of an inductor 14 aand an input terminal of a buffer circuit 62 are connected to a drainterminal of the transistor 61, and one end of the other inductor 14 b isconnected to an output terminal of the buffer circuit 62.

The buffer circuit 62 is constituted of a circuit formed on anintegrated circuit, e.g., a common source circuit or a source followercircuit, and its gain is varied. By changing a gain of the buffercircuit 62, a distribution ratio of signal levels to the inductors 14 a,14 b is changed as in the cases of the aforementioned embodiments.Accordingly, mutual inductance is changed to enable a change ininductance between variable inductor terminals 15, 16.

According to the embodiment, the FET is used for the transistor 61 inthe distributor 12. However, as in the cases of the aforementionedembodiments, a bipolar transistor may be used.

Seventh Embodiment

The embodiments have been described by way of case where thedistributors 12 are all in single-end circuitry. However, thedistributor 12 may be constituted of a differential circuit. As avariable inductor of a seventh embodiment of the present invention, anexample where the distributor 12 is constituted of a differentialcircuit is shown in FIG. 9. In the circuit shown in FIG. 9, adifferential amplifier cascode-connected to the distributor 12 is used.

Gate terminals of differential pair transistors 71 a, 71 b and gateterminals of differential pair transistors 72 a, 72 b are connected todifferential input terminals 11 a, 11 b. A source terminal of thedifferential pair transistors 71 a, 71 b is connected to a currentsource 78, and drain terminals of the differential pair transistors 72a, 72 b are connected to a current source 79. Drain terminals of thedifferential pair transistors 71 a, 71 b are connected to sourceterminals of transistors 73 a, 73 b, and the drain terminals of thedifferential pair transistors 72 a, 72 b are connected to sourceterminals of transistors 74 a, 74 b and source terminals of transistors75 a, 75 b.

Gate terminals of the transistors 73 a, 73 b, 74 a, 74 b, 75 a, 75 b areconnected to a control signal input terminal. Drain terminals of thetransistors 73 a, 73 b are connected to differential output terminals 22a, 22 b, and to ends of inductors 76 a, 76 b. Drain terminals of thetransistor 74 a and the transistor 75 b are connected in common to oneend of an inductor 77 a. Drain terminals of the transistor 74 b and thetransistor 75 a are connected in common to one end of an inductor 77 b.The other ends of the inductors 76 a, 76 b, 77 a, 77 b are connected toa power source Vdd which is a constant potential point.

Signals Input+, Input− opposite to each other in phase are input to thedifferential input terminals 11 a, 11 b. The entry of the signalsInput+, Input− opposite to each other in phase is equivalent to reversalof codes of mutual inductance M12 between the inductors 76 a, 77 a andmutual inductance M13 between the inductors 76 b, 77 b. Accordingly, bycontrol signals φ+ and φ−, interconnection for mutual inductance M12 andM13 can be controlled in magnetic flux reinforcing direction and in amagnetic flux weakening direction. Thus, the inductor can be changed.

FIG. 10 shows a distributor provided with a differential circuitaccording to a modification of the circuit configuration shown in FIG.9, in which bipolar transistors are used instead of the FETs. Thedistributor shown in FIG. 10 can be operated in a same manner as that ofFIG. 9, even if the different type transistors are incorporated in thedistributor.

As described above, according to the variable inductor of the embodimentof the present invention, it is possible to electro-magnetically changethe inductor by using the distributor to be formed on the integratedcircuit. Moreover, such a variable inductor is suited for integratedcircuit formation because of good electric characteristics, easyminiaturization and easy achievement of low costs.

<Oscillator Comprising Variable Inductor and Radio Terminal Comprisingthis Oscillator>

Next, description will be made of an oscillator which incorporates theaforementioned variable inductor of the present invention, and a radioterminal which comprises this oscillator.

FIG. 11 is a block diagram schematically showing an oscillator accordingto an embodiment of the present invention.

The oscillator shown in FIG. 11 comprises a core circuit section (VCOcore) 102 which includes an LC resonant circuit having an inductor L0,and an oscillation frequency control section 104-1 (frequencycontroller) which has an inductor L11 to be electro-magnetically coupledto the inductor L0 based on a coupling coefficient k1 and can control acurrent supplied to the inductor L11.

A control signal Control_1 is input to the oscillation frequency controlsection 104-1 and, in accordance with this control signal Control_1, atleast one or both of an amplitude and a phase of a current flowingthrough the inductor L11 is changed. As a result, inductance of theinductor L0 of the core circuit section 102 coupled to the inductor L11is changed to cause a change in an oscillation frequency. For example,when a current of a direction where a magnetic flux is generated in areinforcing direction with a magnetic flux generated in the inductor L0flows through the inductor L11, a value of inductance of the inductor L0becomes large, and an oscillation frequency becomes small in proportionto −½ square of the value of the inductance. When a current of adirection where a magnetic flux generated in the inductor L0 is weakenedflows through the inductor L11, a value of inductance of the inductor L0becomes small and, as a result, an oscillation frequency becomes large.By setting the currents flowing through the inductor L0 and the inductorL11 in the same direction to control current amplitudes, anincrease/decrease of the inductor L0 can be controlled.

Generally, a control range of a frequency obtained by varying a capacityC is about C max/C min=2. On the other hand, in the system of changinginductance, for example, if currents flowing through the inductor L andthe inductor L11 are set to equal values, and k=about 0.7 is set, Lmin=(1−k)L, L max=(1+k) are established to realize a large fluctuationrange of L max/L min=about 6.

Normally, to change an oscillation frequency by a variable capacity,only a change in a range of 5 to 10% of an oscillation frequency isobtained. However, in the oscillator, which comprises the variableinductor as shown in FIG. 11, an oscillation frequency can be changed toa range of 50% to 100%.

The number of inductors L11 is not limited to one. As shown in FIG. 11,a plurality of inductors L11 to L1 n, e.g., inductors L11, may bedisposed. In the circuit shown in FIG. 11, n pieces of oscillationfrequency control sections 104-n are disposed, and inductors L11 to L1 nof the oscillation frequency control sections are electro-magneticallycoupled to the inductor L of the VCO circuit 102 based on couplingcoefficients (k1 to kn).

If it is only one inductor L11 that is electro-magnetically coupled toone oscillation frequency control section 104-1, i.e., the inductor L ofthe core circuit section 102, by changing an amplitude or a phase of acurrent flowing through the inductor L11, it is possible to change aninductance value of the inductor L.

In the circuit which comprises the plurality of inductors L11 to L1 n,the VCO circuit 102 can be controlled on various modes. For example,even if a coupling coefficient is equal to that of the inductor L11 anda current identical to that which flows through the inductor L11 flowsthrough the inductor L1 n, the number of inductors L1 n through whichthe current is turned ON/OFF to flow may be changed to change theactivated inductor L1 n electro-magnetically coupled to the inductor L.

The inductors L11 to L1 n having different coupling coefficients k1 tokn can be prepared to be switched to be used. Further, currents flowingthrough the individual inductors L11 to L1 n are changed to enable moremeticulous control of the core circuit section 102.

FIG. 12 shows a circuit example where an oscillation signal (voltagesignal) is input from a voltage control oscillation circuit 106 of adifferential LC resonant type equivalent to the VCO core circuit 102 tothe oscillation frequency control circuit 104, and this oscillationsignal is subjected to voltage-current conversion to be supplied to theinductors L1, L2 coupled to the inductors L01, L02 of the core circuitsection 102.

In the circuit shown in FIG. 12, one end of the inductor L01 isconnected to the current source having a power supply voltage Vdd, andthe other end is connected to a variable capacitor VC1 which comprises adrain and a diode of a MOS transistor T1. A source of this MOStransistor T1 is grounded.

Similarly, one end of the inductor L02 is connected to the currentsource having the power supply voltage Vdd, and the other end isconnected to a variable capacitor VC2 which comprises a drain and adiode of a MOS transistor T2. A source of this MOS transistor T2 isgrounded.

The drain of the MOS transistor T1 is connected to a gate of the MOStransistor T2, and similarly a drain of the MOS transistor T2 isconnected to a gate of the MOS transistor T1.

A capacity control voltage Vctr1 is supplied to the variable capacitorsVC1, VC2 to decide capacities thereof. A resonance frequency isdetermined in accordance with parallel connections (L1-VC1), (L20VC2) ofthe variable capacitors VC1, VC2 and the inductors L1, L2.

In FIG. 12, the variable capacitors VC1, VC2 are used. However, fixedcapacity capacitors may be used. In the case of using the fixed capacitycapacitors, only a fluctuation portion of frequency control dependent onthe variable capacitors VC1, VC2 is eliminated, and there are no changesin operation.

From the oscillator 106, an output 1 (Output_1) from the drain of theMOS transistor T1 and an output 2 (Output_2) from the drain of the MOStransistor T2 are outputted.

In order to change an oscillation frequency of the output signal, theoutput 1 and the output 2 (Output_1, Output_2) are input to acurrent-voltage conversion circuit (V-I converter) 108 of theoscillation frequency control circuit 104 to control currents flowingthrough the inductor L1 electro-magnetically coupled to the inductor L01based on a coupling coefficient k1 and through the inductor L2electro-magnetically coupled to the inductor L02 based on a couplingcoefficient k2.

Here, it is assumed that k1=0.7, k2=0.7 are set, inductance values ofthe inductors (L01, L02, L1, L2) are equal at L, and currents of equalsizes flow through the inductors.

If a current between the electro-magnetically coupled inductors is in amagnetic field reinforcing direction, an inductance value of theinductor L01 is increased from L0 to 1.7L0.

If a current flows in a magnetic field weakening direction, aninductance value of the inductor L01 is reduced to 0.3L0.

Accordingly, it can be understood that if a lower part of an oscillationfrequency is near fLo=2 GHz, a higher part thereof becomesfHi=(0.3/1.7)−½·fLo, and thus an oscillation frequency of 4 GHz orhigher can be obtained.

FIG. 13 shows a specific example where the voltage-current conversioncircuit 108 is constituted of cascode-connected transistors. A voltagecontrol oscillation circuit 16 of a differential LC resonant type issimilar to the circuitry shown in FIG. 12, and denoted by the samereference numeral in FIG. 13, and thus description thereof will beomitted. Hereinafter, a frequency control section 104 will be described.

A drain of a transistor T12 is connected to the inductor L1, and asource of the transistor T12 is connected to a drain of a transistor T11to which gate an output (Output_1) is input. A source of the transistorT12 is grounded through a current source 110. A source of a transistorT13 which drain is connected to the inductor L2 is connected to thedrain of the transistor T11. A control signal φ+ is input to a gate ofthe transistor T12, and a control signal φ− is input to a gate of thetransistor T13. That is, the inductor L1 is connected to the transistorT12 cascode-connected to the transistor T11 where the source is groundedthrough the current source and the output 1 (Output_1) is supplied tothe gate, and the control signal φ+ is supplied to the gate of thetransistor T12. The inductor L2 is connected to the transistor T13cascode-connected to the transistor T11, and the control signal φ− issupplied to the gate of the transistor T13.

Similarly, a drain of a transistor T15 is connected to the inductor L2,and a source of the transistor T15 is connected to a drain of atransistor T14 to which gate an output (Output_2) is input. A source ofthe transistor T14 is grounded through a current source. A source of atransistor T16 which drain is connected to the inductor L1 is connectedto the drain of the transistor T14. A control signal φ+ is input to agate of the transistor T15, and a control signal φ− is input to thetransistor T16. That is, the inductor L2 is connected to the transistorT15 cascode-connected to the transistor T14 where the source is groundedthrough the current source and the output 2 (Output_2) is supplied tothe gate, and the control signal φ+ is supplied to the gate of thetransistor T15. The inductor L1 is connected to the transistor T16cascode-connected to the transistor T14, and the control signal φ− issupplied to the gate of the transistor T16.

By changing the control signals φ+, φ−, for example, directions ofcurrents flowing through the inductors L1, L2 can be reversed.Additionally, by properly setting potentials of the control signals φ+,φ−, amplitudes of the currents flowing through the inductors L1, L2 canbe changed.

Thus, by controlling the control signals φ+, φ−, the currents flowingthrough the inductors L1, L2 are controlled and, as a result, it ispossible to control inductance values of the inductors L01, L02 coupledto the inductors L1, L2. For example, when a current flowing through theinductor L01 (L02) and a current flowing through the inductor L1 (L2)are in-phase, an inductance value of the inductor L01 (L02) becomeslarge and, as a result, an oscillation frequency becomes low. When acurrent flowing through the inductor L01 (L02) and a current flowingthrough the inductor L1 (L2) are reverse in phase, an inductance valueof the inductor L01 (L02) becomes small and, as a result, an oscillationfrequency becomes high.

FIG. 14 shows an oscillator of an embodiment where a plurality ofdifferential pairs are arranged.

One end of an inductor L01 is connected to a current source having apower supply voltage Vdd, and the other end is connected to a drain of aMOS transistor T1 and a capacitor Cl. A source of this MOS transistor T1is grounded through a current source I0.

Similarly, one end of an inductor L02 is connected to a current sourcehaving a power supply voltage Vdd, and the other end is connected to adrain of a MOS transistor T2 and a capacitor C2 constituted of a drain.A source of this transistor T2 is grounded through the current sourceI0.

The drain of the transistor T1 is connected to a gate of the transistorT2 and, similarly, the drain of the transistor T2 is connected to a gateof the transistor T1.

In the circuit shown in FIG. 14, capacitances of the capacitors C1, C2are fixed. However, variable capacitors may be used as in the case shownin FIG. 12.

In the circuit shown in FIG. 14, an oscillation frequency controlsection is constituted of a plurality of differential pairs. An inductorL1 n one end of which is connected to the power supply voltage Vdd,e.g., an inductor L11, is electro-magnetically connected to an inductorL01 of a VCO circuit 106 based on a coupling coefficient kin, e.g., acoupling coefficient k11, and the other end is connected to a MOStransistor T1 n, e.g., a drain of a transistor T11. A capacitor C1(n),e.g., a capacitor C11, is branched to be connected between a drain ofthe transistor T1 n and the inductor L1 n. An output 1 (Output_1) of theVCO circuit is supplied to a gate of the transistor T1 n, and a sourceof the transistor T1 n is grounded through a variable current source In,e.g., a variable current source I1.

Similarly, an inductor L2 n one end of which is connected to the powersupply voltage Vdd, e.g., an inductor L21, is electro-magneticallycoupled to an inductor L02 of the VCO circuit based on a couplingcoefficient k2 n, e.g., a coupling coefficient k21, and the other end isconnected to a MOS transistor T2 n, e.g., a drain of a transistor T21. Acapacitor C2(n), e.g., a capacitor C21, is branched to be connectedbetween a drain of the transistor T2 n and the inductor L2 n.

An output 2 (Output_2) of the VCO circuit 106 is supplied to a gate ofthe transistor T2 n, and a source of the transistor T2 n is groundedthrough the variable current source In.

For the capacitor C1 n and the capacitor C2 n, e.g., the capacitor C11and the capacitor C21, the other ends connected to the inductors areinterconnected.

A plurality of such differential pairs are prepared, and current valuesof variable current sources I0 to In are changed or turned ON/OFF tochange inductance of the inductors L01, L02 of the VCO circuit 106.Accordingly, it is possible to control an oscillation frequency.

FIG. 15 shows an oscillator of an embodiment which uses a variable phaseshifter and a variable gain amplifier.

A VCO circuit 106 has components similar to those shown in FIG. 12,denoted by similar reference numerals, and description thereof will beomitted. Buffer circuits 112, 113 having large impedance seen from theoutside are connected to an output terminal.

Output signals (Output_1, Output_2) from the oscillator are input intovariable phase shifters 116, 118 of an oscillation frequency controlcircuit 104. The phases of the input signals are properly controlled byvariable phase shifters 120, 122 and are output from the shifters120,122. Subsequently, a current is controlled through variable gainamplifiers 124, 126, and a current which phase/current amplitude valueis controlled is supplied to an inductor L1 electro-magnetically coupledto an inductor L01 based on a coupling coefficient k1.

Similarly, a current flowing through an inductor L2 coupled to aninductor L02 based on a coupling coefficient k2 is controlled.

By properly controlling the phase shifting and the current values,inductance of the inductors L01, L02 of the VCO circuit 106 can bevaried, whereby an oscillation frequency can be changed and controlled.

The circuit example has been described by referring to the differentialtype. However, it can be similarly applied in the case of a singlephase. Description will be made of a circuit example where a variableinductor is applied to Colpitts oscillation circuit by referring to FIG.16.

The other end of an inductor L one end of which is connected to a powersupply voltage Vdd is connected to a drain of a MOS transistor T, and asource of the transistor T is grounded through a resistor R. An inductorL0 and the transistor T are branched from a connection point, andcapacitors C1, C2 are connected in series to be grounded. A source of atransistor T1 is connected to a connection point between the capacitorC1 and the capacitor C2.

An output from the connection point between the inductor L0 and thecapacitor C1 is input to a frequency control section 128 (frequencycontrol), and connected to the power supply voltage Vdd to control acurrent flowing through an inductor L1 electro-magnetically coupled tothe inductor L0 based on a coupling coefficient k. Accordingly, it ispossible to change and control an oscillation frequency.

The embodiment has been described based on the use of the MOStransistor. Needless to say, however, a circuit having a similarfunction can also be realized by a bipolar transistor or the like.

The inductor constituting the aforementioned oscillator of the inductorvariable type, i.e., the inductor L1 electro-magnetically coupled to theinductor L0 of the LC resonant circuit side based on the couplingcoefficient k, can be realized by various constitutions. For example, asshown in FIG. 17, a constitution can be employed where a spiralconductor constituting the inductor L and a spiral conductorconstituting the inductor L1 are arrange to be symmetrical. In thiscase, the conductors intersect each other at a center of FIG. 17, andthis intersection portion is constituted so as to overlap the conductorsthrough an insulating film. Accordingly, portions other than theintersection area can be realized by one wiring layer.

When a plurality of inductors L11 . . . L1 n electro-magneticallycoupled to the inductor L01 as shown in FIG. 18, a constitution can beemployed where the inductors L11 . . . L1 b of one turn are arranged inan inner peripheral side of the inductor L01 of one turn.

It is not necessary to arrange conductor patterns on the same plane. Forexample, as shown in FIG. 19, a multilayer wiring can be used tolaminate the inductors L01, L11 . . . L1 n. In FIG. 19, there is aninsulating layer between coil conductors which constitute the inductorsL01, L11 . . . L1 n. However, in FIG. 19, the insulating layer isomitted to simplify the drawing.

Thus, by using the oscillator of a wide variable frequency range, it ispossible to miniaturize the radio terminal which is suited to aplurality of communication systems having different frequencies. Thatis, in the conventional radio terminal, in order to supply anoscillation frequency corresponding to each communication system, anoscillator corresponding to each oscillation frequency must be mounted.However, according to the oscillator of the embodiment of the presentinvention, since the variable frequency range is wide, the communicationsystems which use various oscillation frequencies can be deal with byone oscillator. FIG. 20 shows a block of a radio terminal which supportsvarious communication systems which use communication frequencies of 2GHz, 2.4 GHz, 5 GHz. Signals of the communication systems are suppliedto intermediate frequency sections (IF) 136-1, 136-2, 136-3 throughantennas 130-1, 130-2, 130-3, low noise amplifiers 132-1, 132-2, 132-3,and mixers (MIX) 134-1, 134-2, 134-3. Outputs from the intermediatefrequency sections (IF) 136-1, 136-2, 136-3 are supplied to a base bandprocessing section. Signals of oscillation frequencies corresponding tocommunication systems are supplied to the mixers (MIX) 134-1, 134-2,134-3. In this circuit example, oscillation frequencies of 1. xGHz, 2.xGH, 4. xGHz are supplied from the oscillator 138. For the oscillator138 which supplies the oscillation frequencies, the oscillator of theaforementioned embodiment of the present invention is used. A signal ofa desired frequency obtained by changing inductance is switched by aswitch to be supplied to the mixer MIX of each communication system.

As described above, according to the radio terminal of the embodiment ofthe present invention, without disposing a plurality of oscillators, aplurality of communication systems having different frequency bands,e.g., a potable telephone (PDC, W-CDMA), a Bluetooth®, a radio LAN (2.4GHz, 5 GHz) etc., can be supported by one radio terminal. According tothe oscillator of the embodiment of the invention, a wide variablefrequency range can be realized and, as a result, oscillationfrequencies can be supplied to the plurality of communication systems byone oscillator.

<Amplifier Comprising Variable Inductor and Radio Terminal Having thisAmplifier>

Next, description will be made of an amplifier which comprises thevariable inductor of the embodiment of the present invention.

FIG. 21 is a block diagram schematically showing an amplifier accordingto an embodiment of the present invention.

The amplifier shown in FIG. 21 comprises an amplifier circuit 140 whichincludes an inductor La for degeneration, and a control section 142which comprises an inductor Lc electro-magnetically coupled to theinductor L based on a coupling coefficient k.

The control section 142 receives an input signal Input_1 to theamplifier circuit 140, and supplies a signal current to the inductor Lc.An output Output_1 of the amplifier is changed in characteristics inaccordance with an inductance value of the inductor La changed by thesignal current supplied to the inductor Lc.

In the circuit shown in FIG. 21, only one inductor Lc is shown. However,the number is not limited to one, and a plurality of inductors Lc-1 toLc-n may be disposed. If the plurality of inductors Lc-1 to Lc-n aredisposed, various controls can be realized. For example, an arrangementcan be made where inductors Lc-n having equal coupling coefficients andequal flowing currents are prepared, these currents are turned ON/OFF tochange the number of inductors Lc-1 to Lc-n through which the currentsflow, and thereby the activated inductors Lc-n electro-magneticallycoupled to the inductor L are changed.

Additionally, inductors Lc-1 to Lc-n having different couplingcoefficients k1 to kn are prepared to be used switchingly. Further, moremeticulous control can be carried out by changing currents which flowthrough the inductors Lc-1 to Lc-n.

Inductance of the inductor La can be seemingly changed by changing aphase as in the case of control of the amount of a current includingON/OFF control.

For simpler control, a system may be employed where one inductor Lc isprepared, and a current flowing through this inductor is subjected tobinary control of ON/OFF. In this control system, satisfactory effectsof amplification characteristic varying can be exhibited.

A specific example of a differential amplifier will be described byreferring to FIG. 22.

As shown in FIG. 22, a source of a MOS transistor M1 to which gate aninput signal Input_1 is input is connected through a degenerationinductor L1 to a current source I1. The other end of this current sourceI1 is grounded. An inductor 13 is connected to a drain of the MOStransistor M1, and the other end of the inductor L3 is connected to apower supply potential Vdd.

An input signal Input_2 is input to a gate of a MOS transistor M2 whichconstitutes a differential pair. As in the case of connections of theinductor L1, the MOS transistor M1 and the inductor L3, a degenerationinductor L2 is connected to a source of the MOS transistor M2, andinductor L4 is connected to a drain. The inductor L2 is connected to thecurrent source I1, and the inductor L4 is connected to power supplypotential Vdd.

Output signals Output_1, Output_2 are outputted from the drains of theMOS transistors M1, M2. The control section comprises MOS transistors M3and M4 to which gates input signals Input_1, Input_2 of the differentialamplifier are input. An inductor L5 connected to the inductor L12 basedon a coupling coefficient k1 is connected to a source of the MOStransistor M3, and grounded through the variable current source 12.Similarly, a source of the MOS transistor M4 is connected to thevariable current source 12 through an inductor L6 coupled to theinductor L2 based on a coupling coefficient k2. Drains of the MOStransistors M3, M4 are connected to the power supply potential Vdd.

In order to change characteristics of an output signal from thedifferential amplifier, e.g., distortion characteristics or the like, acurrent value of the variable current source 12 is changed. It isassumed that a current change of the variable current source 12 is abinary of a current ON/OFF. When the variable current source 12 is ON,it is assumed that a pair of electro-magnetically coupled inductors L1,L5 and a pair of electro-magnetically coupled inductors L2, L6 arearranged in magnetic field weakening directions. In such a circuit, whenthe variable current source 12 is ON, inductance of the degenerationinductor looks small, and thus low noise can be achieved by a high gain.That is, a high-gain/low-noise mode can be set. When the variablecurrent source 12 is OFF, since no magnetic fields are generated tocancel magnetic fields generated in the inductors L1, L2, compared withthe case where the variable current source 12 is ON, degenerationinductance looks large. In this case, good distortion characteristicscan be obtained. That is, a mode which attaches importance to distortioncharacteristics can be set.

Therefore, when a differential amplifier is constituted, it is onlynecessary to design a degeneration inductor so that it can haverelatively large inductance.

In the foregoing, the current value of the variable current source 2 iscontrolled by a binary of ON/OFF. However, desired amplifiercharacteristics can be obtained by the current value in steps orcontinuously.

FIG. 23 is a block diagram schematically showing a differentialamplifier according to another embodiment of the present invention.

The amplifier shown in FIG. 23 comprises a first amplifier circuit 150which includes a degeneration inductor La, and a second amplifier 152which includes an inductor Lc electro-magnetically coupled to theinductor La based on a coupling coefficient k.

The second amplifier 152 to which the same input signal Input_1 as thatof the first amplifier 150 is entered functions as a control section tocontrol amplifier characteristics of the first amplifier 150, e.g.,distortion characteristics or the like. An output of the secondamplifier 152 is added to an output of the first amplifier 150 to beoutputted as an output signal Output_1.

As in the case of the aforementioned embodiment, the second amplifier152 equivalent to the control section receives an input signal Input_1to the amplifier circuit 150, and supplies a signal current to theinductor Lc. The output Output_1 of the first amplifier 150 is changedin characteristics in accordance with an inductance value of theinductor L changed by the signal current supplied to the inductor Lc. Atthis time, the output of the second amplifier 152 is returned to theoutput Output_1 so that an amplification rate is increased with respectto the input signal Input_1.

FIG. 24 shows a specific circuit example of the differential amplifiershown in FIG. 23.

A source of a MOS transistor M1 to which gate an input signal Input_1 isconnected to a current source I1 through a degeneration inductor L1. Theother end of the current source Ii is grounded. An inductor L3 isconnected to a drain of the MOS transistor M1, and the other end of theinductor L3 is connected to a power supply potential Vdd.

An input signal Input_2 is input to a gate of the MOS transistor M2which constitutes a differential pair. As in the case of connections ofthe inductor L1, the transistor M1 and the inductor L3, a degenerationinductor L2 is connected to a source of the MOS transistor M2, aninductor L4 is connected to a drain thereof, the inductor L2 isconnected to the current source I1, and the inductor L4 is connected tothe power supply potential Vdd.

Output signals Output_1, Output_2 are outputted from the drains of theMOS transistors M1, M2. The second amplifier 152 equivalent to thecontrol section comprises a MOS transistor M3 and a MOS transistor M4 towhich gates the input signals Input_1, Input_2 of the differentialamplifier are input.

An inductor L5 coupled to the inductor L1 based on a couplingcoefficient k1 is connected to a source of the MOS transistor M3, andgrounded through a variable current source I2. Similarly, a source ofthe MOS transistor M4 is connected to the variable current source I2through an inductor L6 coupled to the inductor L2 based on a couplingcoefficient k2.

A drain of the MOS transistor M3 is connected to the drain of the MOStransistor M1, and a current from the variable current source I2 whichflows through the MOS transistor M3 is added to the output signalOutput_1. Similarly, a drain of the MOS transistor M4 is connected tothe drain of the MOS transistor M2, and a current from the variablecurrent source I2 which flows through the MOS transistor M4 is added tothe output signal Output_2.

By employing such circuitry, the current flowing through the controlsection can be used to increase current utilization efficiency for theamplifier as a whole.

Though depending on designs, for example, if a current flowing through anormal amplifier is 5 mA, it may be necessary to supply a current up to10 mA in order to realize low distortion characteristics. On the otherhand, in the circuit shown in FIG. 24, I1=12=2.5 mA is set whenhigh-gain/low noise is achieved, characteristics similar to those when acurrent of 5 mA is supplied in the normal amplifier can be expected. Onthe other hand, when low distortion characteristics are achieved, onlyby turning OFF the current source I2 to supply a current of 2.5 mA tothe current source I1, it is possible to realize low distortioncharacteristics by a large degeneration inductor effect.

In the aforementioned circuit, amplifier characteristics can be varied,which is accompanied by a change in input impedance of the amplifier.Generally, the input impedance change of the amplifier is not desirable.Thus, an impedance control section is preferably disposed in the inputsection of the amplifier.

By referring to FIG. 25, description will be made of a circuit whichcomprises a variable resistor as an input impedance control section.

In the circuit shown in FIG. 25, a variable resistor Rv is insertedbetween an input terminal Input_1 and an input terminal Input_2. Thecircuit shown in FIG. 25 has other components excluding the variableresistor Rv which are similar to those of FIG. 24.

The variable resistor Rv may be constituted by using an FET as shown inFIG. 26A, or by combining a fixed resistor with a switch as shown inFIG. 26B.

In the circuit shown in FIG. 25, as shown in FIG. 27, input impedance ofan amplification stage is represented by the following equation (5):Zin=Lgm/C+j(ωL−1/ωC)   (5)

In the equation (5), if a bias current flows, there is a real partdefined by a first item of the equation (5). However, if the biascurrent is OFF, gm=0 is set, and thus the input impedance has no realparts. Therefore, in order to approximate the input impedance when thebias current is turned OFF to the input impedance when the bias currentflows, a real part of the impedance must be compensated for.

To simplify explanation, a resistance value of the variable resistor Rvis set to be binary, i.e., Open/ON. In the circuit shown in FIG. 25, itis assumed that the inductor L1 and the inductor L5, and the inductor L2and the inductor L6 are arranged in magnetic flux canceling directions.In the assumption, if the current source I1 and the current source I2are both ON, total degeneration inductance becomes small to allow anoperation by a high gain/low noise. Here, the variable resistor Rv isassumed to be open. On the other hand, if only the current source I1 isturned ON while the current source I2 is turned OFF, total degenerationinductance becomes large to allow an operation by low distortion.However, since no bias currents follow through the transistor M3 and thetransistor M4, the real part of the impedance disappears. In order tocompensate for the real part of the impedance, a resistance value of thevariable Rv is turned ON to set a proper resistance value in thevariable resistor Rv, whereby it is possible to realize input impedanceclose to that when an operation is carried out on a high-gain/low-noisemode.

If current control by the variable current source I2 is carried out insteps or continuously in a binary or more values, the resistance valueof the variable resistor Rv may also be controlled in step or continuouschanges. In such a case, it is only necessary to change a voltage φapplied to the gate by a type which uses an FET similar to that shown inFIG. 26A.

FIG. 28 shows circuitry where a gain can be varied even at a common gatecircuit of a rear stage in the circuit shown in FIG. 25.

Common gate MOS transistors M11 a, M11 b are disposed between the MOStransistor M1 and the inductor L3 of FIG. 24, the transistor M11 a isturned ON/OFF by a gate signal φ2, and a gate of the transistor M11 b isconnected to, e.g., a power source, to be always ON.

A MOS transistor M12 is arranged so that the drain of the MOS transistorM1 can bypass the inductor L3 and the MOS transistor M11 a to beconnected to a power supply potential Vdd. This MOS transistor M12 isdriven by a gate signal φ1.

Similarly, MOS transistors M21 a, M21 b are disposed between the MOStransistor M2 and the inductor L4, the transistor M21 a is turned ON/OFFby a gate signal 42, and the transistor M21 b is always ON. A MOStransistor M22 is arranged so that the drain of the MOS transistor M2can bypass the inductor L4 and the MOS transistor M21 to be connected tothe power supply potential Vdd. This MOS transistor M22 is driven by agate signal φ1.

It is assumed that the transistors M11 a, M12 and the transistors M21,M22 are constituted of MOS transistors of equal sizes, and thetransistor M11 b and the transistor M21 b are constituted of MOStransistors of small sizes. When the gate signal 41 is turned OFF andthe gate signal 42 is turned ON, signals supplied through the transistorM11 a, the transistor M11 b, the transistor M21 a, the transistor M21 bare outputted. On the other hand, when the gate signal 41 is turned ONand the gate signal 42 is turned OFF, only a signal supplied through thetransistor M11 b, the transistor M21 b is outputted, and the othersignal is discarded through the transistor M12, the transistor M22.Consequently, a gain is reduced. By using a gain switching operation andthe gain switching to change the degeneration inductor at the same time,it is possible to realize a larger gain switching width.

FIG. 29 shows a circuit example where an inductance variable circuit isused to adjust input impedance.

In place of the variable resistor Rv shown in the circuit of FIG. 25, animpedance adjustment circuit 160 is disposed which has componentssimilar to the transistor M2 and the inductor L1, the transistor M2 andthe inductor L2, the transistor M3 and the inductor L5, and thetransistor M4 and the inductor L6. That is, a drain of the MOStransistor M5 is connected to the power supply potential Vdd, a sourcethereof is connected to a current source I3 through an inductor L7, andan input signal Input_1 is supplied to a gate thereof.

A drain of the MOS transistor M6 which constitutes a differential pairwith the MOS transistor M5 is connected to the power supply potentialVdd, a source thereof is connected to the current source I3 through aninductor L8, and an input signal Input_2 is supplied to a gate thereof.

As a circuit to control inductance of the inductors L7, L8, MOStransistors M7 and M8 are disposed, to which gates input signalsInput_1, Input_2 of a differential amplifier 162 are input. An inductorL9 which has a coupling coefficient k3 with the inductor L7 is connectedto a source of the MOS transistor M7, and grounded through a variablecurrent source I4. Similarly, a source of the MOS transistor M8 isconnected to the variable current source I4 through an inductor L10which has a coupling coefficient K4 with the inductor L8. Drains of theMOS transistors M7, M8 are connected to the power supply potential Vdd.

To simplify explanation, the inductors L1 to L10 are set to equalinductance values, and coupling coefficients K1=K3=K2=K4 are set. Thevariable current sources I2, I4 are changed in a binary of a currentON/OFF, and the coupled inductors are coupled to mutually weakenmagnetic fields.

When the variable current source I2 is turned ON, inductance values ofthe degeneration inductors L1 and L2 look small, and operated on ahigh-gain/low-noise mode. In this case, if the variable current source14 is kept OFF, the impedance adjustment circuit is operated on a pseudolow-distortion mode, and total input impedance is set to a value where ahigh-gain/low-noise mode circuit and a low-distortion mode circuit areconnected in parallel.

On the other hand, when the variable current source I2 is turned OFF,inductance values of the degeneration inductors L1 and L2 look large,and operated on a low-distortion mode. In this case, if the variablecurrent source 14 is kept ON, the impedance adjustment circuit isoperated on a pseudo high-gain/low-noise mode, and total input impedanceis set to a value where a high-gain/low-noise mode circuit and alow-distortion mode circuit are connected in parallel. Thus, not changesoccur even if the operation mode is switched.

Further, the embodiment has been described by way of the circuit exampleof the differential pair. However, similar effects can be obtained evenby a single-end circuit. FIG. 30 is a circuit diagram showing anembodiment applied to a single-phase amplifier circuit.

As shown in FIG. 30, a source of a MOS transistor M1 to which gate aninput signal Input_1 is input is connected to a current source I2through a degeneration inductor L. The other end of the current sourceI1 is grounded. An inductor L3 is connected to a drain of the MOStransistor M1, and the other end of the inductor L3 is connected to apower supply potential Vdd. An output signal Output_1 is outputted fromthe drain of this MOS transistor M1.

A control section comprises a MOS transistor M2 to which gate the inputsignal Input_1 is input. An inductor LC having a coupling coefficient kwith the inductor L is connected to a source of the MOS transistor M2,and grounded through a variable current source I2. A drain of a MOStransistor M3 is connected to the drain of the MOS transistor M1, and acurrent from the variable current source I2 which flows through the MOStransistor M2 is added to the output signal Output_1.

A power source of the variable current source I2 is subjected to, e.g.,ON/OFF control, whereby an inductance value of the degeneration inductorL can be controlled.

A control inductor electro-magnetically coupled to the aforementionedinductor based on a coupling coefficient k, e.g., the inductors L1, L2shown in FIG. 21, can be realized by various constitutions. For example,as shown in FIG. 17, the spiral conductor which constitutes thedegeneration inductor L and the spiral conductor which constitutes thecontrol inductor LC can be arranged to be symmetrical. In this case, theconductors intersect each other at the center in the drawing, and thisportion is formed through an insulating film. Thus, portions other thanthe intersection area can be realized by one wiring layer. The inductoris not limited to the constitution shown in FIG. 17. Any constitutioncan be employed as long as the inductor can be electro-magneticallycoupled, and the inductor may be constituted of the conductor patternshown in FIG. 18 or FIG. 19.

The aforementioned circuit of the embodiment of the present inventionuses the MOS transistors. Needless to say, however, active elements suchas other transistors may be used.

The amplifier of the embodiment of the present invention can be used fora radio communication terminal such as a portable telephone. An exampleis shown in FIG. 31. FIG. 31 is a block diagram of a radio communicationterminal.

An RF input signal from an antenna (ANT) is supplied to an RF signalprocessing section (RF). That is, in the RF signal processing section(RF), the signal is supplied through a switch (T/R) to an RF band-passfilter 1 (RF-BPF1), a low noise amplifier (LNA) and an RF band-passfilter 2 (RF-BPF2), multiplied by a local signal (RF-VCO) at amultiplier (DC), and frequency-converted into an intermediate frequencysignal. This intermediate frequency signal is supplied to anintermediate frequency processing section (IF-Stage) and a base bandsignal processing section (BB-Stage).

A gain control signal (Gain Control) is supplied from a receivedelectric field intensity determination section (RSSI) in the base bandsignal processing section (BB-Stage) to the low noise amplifier (LNA).

A signal to be transmitted is processed reversely to the above. That is,a signal supplied from the base band processing section (BB-Stage) andthe intermediate frequency processing section (IF-Stage) is processed inthe RF signal processing section (RF). In the RF signal processingsection, an intermediate frequency signal is multiplied by a localsignal (RF-VCO) at a multiplier (UC) to be frequency-converted. Theconverted signal is supplied through the RF band-pass filter (BP-BPF) toa power amplifier (PA), and supplied through the switch (T/R) to theantenna (ANT).

The aforementioned amplifier is used for the low noise amplifier (LNA)of such a radio terminal.

For the radio terminal, there are various standards. If a level of an RFsignal to be input is small, amplifier characteristics of amplifying asignal by low noise may be required of the low noise amplifier (LNA). Insuch a case, control is carried out to reduce a degeneration inductancevalue of the LNA. Conversely, if an RF signal is sufficiently large,since it is important to prevent distortion of the RF signal, control iscarried out to increase a degeneration inductance value of the LNA.

Such characteristic control of the low noise amplifier (LNA) can becarried out by using, e.g., a gain control signal (Gain Control). Thecharacteristic control of the amplifier may be carried out based on astandard other than the RSSI.

Thus, by using the amplification characteristic variable amplifier ofthe present invention for the LNA of the RF processing section of theradio terminal, it is possible to adaptively realize desired LNAamplification characteristics without increasing current consumption.

In the amplifier described above with reference to FIGS. 20 to 30, it isassumed that at least pairs of the inductors La, Lc, L2, L5, L2, L6, L7,L9, L8, L10 are interconnected based on coupling coefficients k2, k2, k,k4, and have mutual inductance. However, the amplifier can be realizedeven if the pairs of inductors are not interconnected, and have nomutual inductance. The amplifier, which uses mutual inductance, isadvantageous in that large degeneration can be realized and gooddistortion characteristics can be realized by a smaller current. On theother hand, the amplifier, which comprises the inductors having nomutual inductance, is advantages in that designing becomes easy becauseit is not necessary to use any inductors of complex shapes.

Hereinafter, detailed description will be made of an amplifier whichcomprises inductors having no mutual inductance according to anotherembodiment of the present invention by referring to FIGS. 31 to 38.

The description will be made by way of examples all of which use bipolartransistors. However, the amplifier can be configured by using otheractive elements such as an FET.

FIG. 32 shows basic circuitry of an amplifier according to theembodiment of the present invention. As shown in FIG. 32, a plurality ofamplification stages A1 to An are connected in parallel to an input sideInput and, by selectively operating the amplification stages A1 to An, again switching function is realized. In this circuit, the amplificationstages A1 to An may have same amplification characteristics, differentamplification characteristics or combinations of same amplificationcharacteristic and different amplification characteristics,respectively. When a gain is switched, input impedance is changed.However, this change of the input impedance Zin is compensated for by avariable resistor Rx. That is, irrespective of which one of theamplification stages A1 to An is operated, a value of the variableresistor Rx is set to a proper value so as to prevent a great change inthe input impedance Zin of the entire amplifier. Here, for example, itis assumed that the amplification stage A1 is set as amplification stagehaving a high gain and good distortion characteristics by currentconsumption of a certain level, and the amplification stage A2 is set asan amplification stage having a low gain and good distortioncharacteristics by small current consumption. If a high gain isrequired, the amplification stage A1 is operated. If a low gain isrequired, the amplification stage A2 is operated. Accordingly, a desiredgain and desired distortion characteristics can be achieved, and currentconsumption can be limited to a minimum. By setting the variableresistor Rx to a proper value, it is possible to limit a change small inthe input impedance Zin when the operations of the amplification stagesAl and A2 are switched to change the gain.

FIG. 33 shows a circuit of a first embodiment to realize the variableresistor Rx of the circuit of FIG. 32. The variable resistor Rx whichadjusts input impedance is realized by using fixed resistors R1 to Rm,and switches SW1 to SWm. The switches SW1 to SWm are properly switchedto select the fixed resistors R1 to Rm, whereby a desired resistancevalue is given to the resistor Rx of the input side.

FIG. 34 shows a circuit of a second embodiment to realize the variableresistor Rx of the circuit of FIG. 32. The variable resistor Rx whichadjusts input impedance is constituted of an FET 172. By applying aproper control voltage to a control gate Vctr1 of the FET 172, a desiredresistance value can be obtained.

FIG. 35 shows a circuit of an embodiment to realize the variableresistor Rx of the circuit of FIG. 34. In the circuit shown in FIG. 35,two common emitter amplification stages A1 and A2 are connected inparallel, and a MOSFET is connected to an input stage to adjust inputimpedance. A degeneration inductor L1 of the amplification stage A1 hasa small inductance value so as to achieve a high gain. Additionally, adegeneration inductor L2 of the amplification stage A2 has a largeinductance value to achieve a low gain and good distortioncharacteristics. FIG. 36 shows a simulation result of input impedance(admittance) of a portion excluding the MOSFET 172 of the input sectionin the circuit shown in FIG. 35. FIG. 36 shows reflection coefficientson the admittance in a first case where a circuit constant and anoperation point are properly set, and the amplification stage A1 isturned ON and the amplification stage A2 is turned OFF to achieve a highgain, and in a second case where the amplification stage A1 is turnedOFF and the amplification stage A2 is turned ON to achieve a low gain.As apparent from FIG. 36, proper resistors are connected in parallel,i.e., the MOSFET 172 connected to the input section is turned ON, andaccordingly impedance of the first and second cases shown in FIG. 36 canbe set substantially equal to each other. Current consumption is about 3mA when a high gain is achieved, and current consumption is about 1.5 mAwhen a low gain is achieved. In the case of the low gain, currentconsumption is reduced. In the case of the low gain, since a value ofthe degeneration inductor L2 is large while the current consumption issmall, good distortion characteristics can be achieved.

FIG. 37 shows a circuit of another embodiment to realize the variableresistor Rx of the circuit of FIG. 34. In the circuit shown in FIG. 37,two differential amplification stages A1 and A2 are connected inparallel, and a MOSFET 172 is connected to an input stage to adjustinput impedance. Here, to achieve a high gain, the amplification stagesA1, A2 are both turned ON. When the two amplifiers A1, A2 are operated,the degeneration inductor also looks small, and thus a high gain can beachieved. To achieve a low gain, only the amplification stage A1 isturned ON, and the amplification stage A2 is turned OFF. Thus, sinceonly the inductor L1 among the degeneration inductors can be seen,degeneration becomes large, and good distortion characteristics can beachieved.

FIG. 38 shows a circuit of an embodiment to realize the variableresistor Rx of the circuit of FIG. 33. In the circuit of FIG. 38, serialcircuits of fixed resistors R1 to R3 and switches SW1 to SWm areconnected in parallel between inputs Input_1, Input_2. In this circuit,by selectively switching ON/OFF the switches SW1 to SWm, the resistor Rxof an input side can be properly set. That is, by properly switching theswitches SW1 to SWm, the fixed resistors R1 to Rm are selected to give adesired resistance value to the resistor Rx of the input side.

FIG. 39 shows a circuit example where the amplifier circuit of FIG. 32is applied to a low nose amplifier of a radio terminal. In the circuitof the radio terminal shown in FIG. 39, an RF input signal from anantenna (ANT) is input to an RF signal processing section (RF), andsupplied to the low noise amplifier (LNA) of the RF signal processingsection (RF) described above with reference to FIG. 32. An output signalfrom the low noise amplifier (LNA) is multiplied by a local signal(RF-VCO) at a multiplier (DC) to be frequency-converted into anintermediate frequency signal. This intermediate frequency signal issupplied through a band-pass filter (BPF) to an intermediate frequencyamplifier (IF-AMP) of an intermediate frequency processing section(IF-Stage). An output from the intermediate frequency amplifier (IF-AMP)is supplied through a quadrature demodulator (QDEM) to a base bandsignal processing section (BB-Stage) to be processed.

In the circuit of the radio terminal, the amplifier circuit of FIG. 32is applied to the low noise amplifier to realize gain switching and tomake input impedance constant. By the gain switching function, a dynamicrange of the radio terminal is widened, and the input impedance of thelow noise amplifier is not changed. Accordingly, an alignment can alwaysbe made with desired impedance such as 50 Ω easily by using a singleinput alignment circuit. Moreover, low power consumption is required ofthe circuit used for the radio terminal. However, since the low noiseamplifier, to which the present invention is applied, can be operated bya minimum necessary current, it leads to lower power consumption of theradio terminal.

Additional advantages and modifications will readily occur to thoseskilled in the art. Therefore, the invention in its broader aspects isnot limited to the specific details and representative embodiments shownand described herein. Accordingly, various modifications may be madewithout departing from the spirit or scope of the general inventionconcept as defined by the appended claims and their equivalents.

1-10. (canceled)
 11. An oscillator comprising: a voltage controloscillation circuit having a first inductor; and a frequency controlcircuit having a second inductor electro-magnetically coupled to thefirst inductor, configured to supply a control current to the secondinductor, and controls an oscillation frequency of the voltage controloscillation circuit by changing the control current to change aninductance value of the first inductor.
 12. The oscillator according toclaim 11, wherein the frequency control circuit includes a plurality ofsecond inductors, and controls control currents flowing through theplurality of second inductors.
 13. The oscillator according to claim 11,wherein the frequency control circuit changes at least one of anamplitude value and a phase of the current flowing through each of thesecond inductors.
 14. The oscillator according to claim 11, wherein thefrequency control circuit comprises a plurality of second inductorsconnected to the first inductor based on different couplingcoefficients. 15-25. (canceled)